Device for altering tone quality in electronic musical instrument or the like



p 1. 1964 J. P. WHITE 3,147,334

DEVICE FOR ALTERING TONE QUALITY IN ELECTRONIC MUSICAL INSTRUMENT OR THELIKE Original Filed Sept. 28, 1955 5 Sheets-Sheet 1 FIGI TRANSMISSIONSOLID PHASE SHlFT-DOTTED AUDIO FREQUENCY FIG 2 SPEAKER INPUT X 5ISflM-ATIN6 AMPLIFIER ALL-PASS PHASE smrr 35 CIRCUIT 4 E NETWCRMS) 36 QEQUALIZER GAIN CONTROL INVENTOR: I JAMES PAUL WHITE BY WW ATTYS Sept. 1,1964 J. P. WHITE 3,147,334

DEVICE FOR ALTERING TONE QUALITY IN ELECTRONIC MUSICAL INSTRUMENT OR THELIKE Original Filed Sept. 28, 1955 5 Sheets-Sheet 2 LAST PAIR 7 QITERMEDIATE PAIRS ouTPu'r INVENTOR: v JAMES PAUL WHITE BY 4W ATTYS.

Sept. 1, 1964 MIXER J. P. WHITE DEVICE FOR ALTERING TONE QUALITY INELECTRONIC MUSICAL INSTRUMENT OR THE LIKE Original Filed Sept. 28, 19555 Sheets-Sheet 5 INVENTOR JAMES PAUL WHITE ATTYS.

Sept. 1, 1964 I J. P. WHITE 3,147,334

DEVICE FOR ALTERING TONE QUALITY IN ELECTRONIC MUSICAL INSTRUMENT OR THELIKE 5 Sheets-Sheet 4 Original Filed Sept. 28, 1955 FIG4.

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p 1, 1964 J. P. WHITE DEVICE FOR ALTERING TONE QUALITY IN ELECTRONICMUSICAL INSTRUMENT OR THE LIKE 5 Sheets-Sheet 5 Original Filed Sept. 2a,1955 ELECTRO- MUL'I'I- ELECTRO- Acpus-nc RESONANT ACOUSTIC 05/SIGNAL onACOUSTIC on uTPuT\/09 m ELECTRO- on ELECTRO- MECHANICAL MECHANICALuecmmcm.

TRANSDUCER ocvlcc ramsoucsn //6 4 Y OUTPUT l2 55 \|NPur I FIG] M8 v //9ELECTRO- MULTI- ACOUSTIC RESONANT 20 ACOUSTIC //7/SIGNAL on Acousnc Aeueasv mwur ELECTRO- on omzcnv MECHANICAL MECHANICAL mu-co rnmsoucznDEVICE FIQB,

i INVENTOR JAMES PAUL WH IT E BY WA W ATTYS.

United States Patent DEVICE FOR ALTERING TONE QUALITY IN ELEC- TRONECMUSECAL KNSTRUMENT OR THE LIKE James Paul White, 1441 Manor Lane, RD. 4,

Norristown, Pa.

Original application ept. 28, 1955, Ser. No. 537,134, now Patent No.3,031,909, dated May 1, 1962. Divided and this application Nov. 8, 1961,Ser. No. 150,916

21 Claims. (Cl. 84-125) The present invention relates to electrical andelectronic musical instruments, and more particularly to the improvementof the tone quality of such instruments.

It is frequently observed that in spite of the best efforts of themakers of electronic organs and associated musical instruments, thereyet remains a certain lack in the tonal quality of these instrumentswhich makes them inferior to the actual instruments of the orchestra.The present invention provides a novel method of overcoming this tonaldeficiency of electronic and electrical musical instruments.

It is generally recognized that sound produced by several instrumentsplaying in unison, known as chorus effect, is more pleasing than thesound of a single instrument. Likewise, the sound of a single instrumentis more pleasing than the sound of an electronically generated tone.These effects are due to a preference by the ear for sounds producing athrobbing or beating sensation. The inequalities of pitch existing amongseveral instruments playing in unison produce a strong beatingsensation; the beating among the harmonics of a single instrument (to bedescribed) produces a modest beating effect; and an electronicallygenerated tone produces an almost negligible beating effect.

A single orchestral instrument produces a beating effect for tworeasons: (1) The pitch is not absolutely stable, and (2) The phase shiftbetween the primary source of vibration and the air is not constant, butis a function of frequency; that is, the phase shift through theinstrument (for example, from mouthpiece to the air in the case of atrumpet) would be an undulating (up and down) curve when plotted againstfrequency.

These two facts cooperate to produce a beating effect as follows. All ofthe harmonics of the primary source of vibration are varying infrequency at once by reason of the first factor. As each harmonic sovaries, its phase will change an additional amount dependent on theshape of the phase curve (second factor) in the vicinity of the harmonicfrequency in question. Since this curve will have various positive andnegative slopes at the different harmonic frequencies, the additionalrate of change of phase of the various harmonics will in general beunequal so that (as is well known in the theory of frequency modulation)the frequencies of the harmonics will be shifted instantaneously withrespect to the fundamental frequency in a somewhat random manner. Hence,the instantaneous frequencies of the audible harmonics will not be inexact (integral) harmonic relation, and hence will beat with each otherand produce a sound having a much more pleasing musical effect than thesound produced by an instrument with all harmonics in exact harmonicrelationship at all times, such as an electronically generated tone.

The principles as outlined above for non-electric musical instrumentsare applied in this invention to electric and electronic musicalinstruments, by providing an undulating phase shift vs. frequencycharacteristic (similar to that described) in the transmission pathbetween the source of electrical vibrations and the ear of the listener.

It is a fundamental object of the present invention to provide a devicefor use with electrical and electronic W 3&47334 Patented Sept. 1, 1964musical instruments to improve the tone quality, and produce new andinteresting tone qualities.

It is another object of the invention to provide a device for use withelectrical and electronic musical instruments to alter the tonalcharacter in a manner closely analogous to the manner in which theinitial mechanical vibrations of most non-electronic musical instrumentsare altered before the principal acoustic radiation takes place.

Other and further objects and advantages of the invention will becomeapparent from the following description, taken in conjunction with theaccompanying drawings; in which FIG. 1 illustrates the type ofcharacteristics desirable in a device to alter and/or improve the tonalquality of electrical and/ or electronic musical instruments. FIG. 1also illustrates the characteristics of most non-electric musicalinstruments with regard to the transmission and phase shift from theprimary vibration generator to the surrounding air.

FIG. 2 is essentially a block diagram of an electronic filter embodyingthe principles of the present invention, and showing, symbolically, theaction of the vibrator addition.

FIG. 2a is a Nyquist diagram of the electronic filter, and shows thelocus of certain vectors useful in understanding the operation of theelectronic filter.

FIG. 3 is a schematic wiring diagram of the electronic filter shown inblock diagram in FIG. 2.

FIG. 3a is a schematic wiring diagram of a vibrato addition to thecircuit of FIG. 3.

FIG. 4 is a block diagram of an electroacoustic or electromechanicalembodiment of the principles of the present invention.

FIG. 4a is a combination schematic diagram and mechanical illustrationof a representative physical embodiment of the block diagram of FIG. 4.

FIG. 4b shows a modification of the embodiment of FIG. 4a for thepurpose of producing a vibrato.

FIG. 5 is a block diagram of an electroacoustic or electromechanicalembodiment of the present invention in which means are provided tominimize the effects of reverberation due to the multiresonant acousticor mechanical device.

FiG. 5a is a combination schematic diagram and mechanical illustrationof a representative physical embodiment of the block diagram of FIG. 5.

FIG. 6 is a block diagram of another type of electroacoustic orelectromechanical device embodying the principles of the presentinvention.

FIG. 6a is a combination schematic diagram and mechanical illustrationof a representative physical embodiment of the block diagram of FIG. 6.

FIG. 7 is a block diagram of another embodiment of the present inventionin which acoustic energy is directly radiated without the use of aconventional loudspeaker.

FIG. 8 is a schematic diagram of a representative lumped constantnetwork embodying the principles of the present invention.

In FIG. 1 appears a diagram of the transmission curve 1 and phase shiftcurve 2 typical of the body of stringtype orchestral instruments and ofthe air column of wind-type orchestral instruments and also of thevarious tone quality improvement devices herein disclosed. It has beenshown that the phase shift curve 2 is responsible for the production ofa more pleasing tone quality when a device producing an effect similarto that depicted by FIG. 1 is used in the transmission path between theinitial source of signal and the listener, provided that there is somefrequency instability in the signal applied to the device in question.The use of such devices in conjunction with an electrical or electronicmusical instrument is set forth by the present disclosure.

The difference between the usual tonal control and coloring circuitsfound in all electronic or electric musical instruments and the tonecoloring method discussed here may be summarized as follows. The tonecolor (timbre) circuits generally used in electronic organs have astheir purpose the production of an output wave having the desiredharmonic structure characteristic of each note of the scale for the stopin use. Such tone color circuits as used in electric or electronicorgans will also have a phase shift vs. frequency characteristic, butthe phase shift produced is very much less than that of the methodproposed by the present invention. That is, the total phase change inthe audio band is much greater for the proposed method than for anyexisting electronic musical instrument. By total phase change is meantthe sum of the magnitudes of all phase changes in the audio rangewithout regard to sign; that is, to obtain the total phase change inFIG. 1, we would add the magnitude of the phase change from maximum 67to minimum 68 to the magnitude of the phase change from minimum 68 tomaximum 69, and so on throughout the entire audio frequency range. Thedistinction between the present state of the art and the presentinvention is this: The tone color circuits of electronic organs andsimilar instruments are used to produce relatively gradual changes intransmission with respect to frequency for the purpose of producing anoutput wave of the desired harmonic structure, with only incidentalchanges in phase shift with frequency; on the other hand, the tonecoloring system herein disclosed makes use of rapid changes in phaseshift with respect to frequency for the purpose of ultimately producingbeats in the auditory sensation (as described), with only incidentalchanges in transmission with frequency. In the case of the prior art,the incidental changes in phase shift with frequency are insignificantfor the purpose of ultimately producing an auditory beating sensation,whereas with this invention such changes in phase shift with frequencyare very significant. The phase and amplitude undulations shown in FIG.1 are repersentative only of the type of characteristic desired, and notof the exact number of undulations proposed. In general, the more peaksthere are in these curves, the better the effect. Even if it weredesirable to limit the differences between adjacent maxima and minima ofthe transmission curve 1 of FIG. 1 to say 3 decibels, it would still bepossible to obtain any desired amount of total phase change throughoutthe range by having a sufficiently large number of these relativelysmall transmission peaks.

An electronic device, which may be designated an electronic filter,capable of producing the rapidly varying phase shift with frequency asdesired, is shown in block diagram in FIG. 2, and the actual circuit inFIG. 3. The block diagram, FIG. 2, shows a voltage divider consisting ofresistors 3 and 4. If the rest of the circuit were disconnected, thisvoltage divider would attenuate the input voltage at terminal 12 toproduce the output voltage at terminal 13 by the same amount at allfrequencies within the range of interest. When the rest of the circuitis connected as shown, the impedance from point to ground is no longeralways equal to the resistance 4 but may be either larger or smallerthan resistor 4, depending on the circuit adjustments and the frequency.Hence, the voltage divider consisting of resistor 3 and the effectiveimpedance between point 5 and ground will change its output as thefrequency is varied, so that if the input voltage were constant at allfrequencies, the output voltage would be found to vary in amplitude andphase as the frequency was varied throughout the frequency range ofinterest.

The block diagram of FIG. 2 consists of five parts (in addition toresistor 3). A is an all-pass phase shift network(s); B is an isolatingcircuit, and includes the resistor 4; C is a gain control; D is anequalizer; and E is an amplifier. Although all of these five parts areshown separately, it is nevertheless true that not all are absolutelynecessary for the operation of a circuit of this type. Parts B, C, and Dcould conceivably be omitted (except that the resistor 4 would stillhave to be physically or effectively incorporated into the remainder ofthe circuit), and part E could even be designed to be an integral partof part A. In addition, the relative order of these five parts need notbe as shown in FIGS. 2 and 3, but may be in any convenient sequence aswe proceed around the closed loop from point 5 until we return to point5. Any such changes would not alter the basic operation of the circuit,even though they might make it slightly less flexible. For convenienceand clarity, let us consider the operation in terms of the block diagramof FIG. 2. The five parts of this circuit mentioned constitute afeedback circuit in that the parts are connected I together so as toform a complete closed loop, as shown by the connecting lines andarrows. It is well known to those skilled in the ant that the impedanceto ground looking into such a circuit at a convenient point such aspoint 5, is given by the following formula:

1ABCDE In this formula Zo represents the effective impedance to groundat point 5 with the feedback loop broken at some remote point; the othersymbols represent the transfer function as noted. For example, A in theformula represents the mathematical expression giving the vector outputvoltage of the circuit part A, divided by the vector input voltage ofcircuit part A as a function of frequency, with the circuit part Aactually connected to the rest of the circuit as shown. An examinationof this formula for Z shows that when the quantity -ABCDE in thedenominator approaches minus one, the value of Z will become very large,so that the magnitude of the circuit output voltage at terminal 13 mayapproach that of the circuit input voltage at terminal 12. At otherfrequencies where the quantity -ABCDE does not approach minus one, thevalue of Z will be smaller, so that the circuit output voltage may bemuch smaller than the circuit input voltage. can be made to approach andrecede from the value minus one any desired number of times as thefrequency is varied over the range of interest, the desired number ofpeaks will be introduced into the response vs. frequency curve of theelectrical system to which this circuit is connected, and a large totalphase change will be produced if sutlicient peaks are introduced intothe system.

A diagram showing one possible way in which the quantity ABCDE mightvary with frequency is shown in FIG. 2a. This diagram is a Nyquist typeof diagram, in which the magnitude and phase angle of the quantity-ABCDE are plotted for all frequencies. The curved line is the locus ofthe tip of the vector extending from 0 to 7. The distance on the diagramfrom 0 to 6 represents unit distance, so that at some particularfrequency the quantity ABCDE might be plotted to the same scale as point7, where the distance from 0 to 7 represents the magnitude of thisquantity, and the angle 8 represents its phase angle. As the frequencyvaries, the point 7 will move. The entire locus of point 7 may bedescribed as follows: starting at zero frequency at the point 0, wefollow the arrow clockwise to near the point 9, and continue clockwiseon the approximately circular path through 10 and 9, going around thecenter point 0 as many complete revolutions of 360 degrees as we desirepeaks in the response vs. frequency curve of the electrical system. Itis obvious from the foregoing that each time the point 7 reachesposition 10, a peak will occur, since the point 6 represents theposition of minus one on the diagram. Similarly, each time the point 7reaches position 9 a minimum of transmission through the device vw'lloccur. The ratio of this maximum to minimum output voltage will be thesame as the ratio of the length of the straight line connecting points 6and 9 to that connecting Hence, if the quantity -ABCDE points 6 and 10.After the last revolution, the point 7 will pass near the point 9 andfollow the clockwise path to point 0 which will also correspond toinfiinite frequency. If the frequencies corresponding to the passage ofpoint 7 through point 10 are chosen to be within the frequency band ofinterest, we have succeeded in introducing peaks as desired. It may beseen that the magnitude of the peaks may be controlled by C, the loopgain control, so that the point 10 may be made to fall anywhere on theline segment from 0 to 6. The peaks and the corresponding phase shiftwill be large when 10 falls near 6, and small when 10 is near 0.

The portions of the Nyquist diagram, FIG. 2a, from 0 to near 9 are dueto residual circuit reactances, or they may be controlled by theequalizer, part D. In the description just given of FIG. 2a, the fiveparts of the circuit as shown in FIG. 2 were treated as having themagnitudes of their transfer functions substantially constant over thefrequency range of interest, when the point 7 is travelling around thelarge circular path. The allpass phase shift network(s), A, aredesigned, however, to have a constantly changing phase angle withfrequency throughout the frequency band of interest. It is apparent thatthe Nyquist diagram of FIG. 2a could be varied considerably byadjustment of the Equalizer, D, to have the magnitude of the transferfunction of part D vary with frequency. This would cause the contourtraced by point 7 to differ from that shown, so that the point 7 on itssuccessive revolutions around point 0, would approach the point 6 moreor less closely; that is, there Would be a multiplicity of points 10where the locus intersects the horizontal axis between 0 and 6. Sincethe distance from point 6 to point 10 is proportional to the denominatorof the expression for Z, it is seen that the impedance Z determining themagnitude of the peaks will vary from one revolution of the point 7 tothe next, so that the peaks will have different and controllableheights. That is, each revolution of the point 7 will produce in thiscase a different minimum value of the denominator of Z, and so avariable maximum of Z, and hence a changing peak output as we go throughthe different peaks.

The gain control, C, of FIG. 2, will expand the entire diagram of FIG.2a proportionally about the point 0, so that the point 10 can be made toapproach the point 6 as closely as desired; conversely, the diagram canbe made to contract as desired, until at minimum gain the entire diagramwill coincide with point t), and the output voltage of the circuit willbe constant if the input voltage vs. frequency is constant, therebyeliminating all peaks and phase shift. Hence, if resistor 3'of FIG. 2 islarge compared with the effective impedance at point 5, which is Z, thepeaks may be varied from very large when the points 10 are very nearpoint 6, to very small or vanishing peaks when the points 10 approach orcoincide with the point 0. The phase shift curve will vary in acorresponding manner. To those skilled in the art, it will be obviousthat the phase angle of the denominator of the expression for Z, isshown in FIG. 2a by the angle 11. Therefore, if we assume a zero phaseangle for Z0, the phase angle of Z is merely the negative of the angle11. The magnitude of the variations of the angle 11, formed by thestraight line segments from point 0 to point 6 and from point 6 to point7, will then be the same as the magnitude of the variations of the angleof the impedance Z, and the total phase change of the impedance Z willbe equal to the total angular change of the angle 11. It should be notedthat the total phase change of the output voltage at terminal 13 of FIG.2, with respect to the input voltage at terminal 12, will be somewhatless than the total phase change of the impedance Z, due to the presenceof resistor 3. The larger resistor 3 is made, the more nearly equal willthese two total phase changes become; however, it is impractical to makeresistor 3 too large, because this would reduce the output voltage atterminal 13 to art undesirably small value. In FIG. 2a, it is readilyseen that the maximum value of angle 11 will occur when the straightline segment from point 6 to point 7 is tangent to the circular paththrough point 10 in the second quadrant; and similarly, the minimum willoccur when the point of tangency lies in the third quadrant. It is alsoobvious that the closer the points 10 are to point 6, the larger will bethe variations in the angle 11, and the greater will be the total phasechange in the output voltage. Thus, large variations in the total phasechange may be produced by adjustment of the gain control, C, since thisgain control will control the position of the points 10 between thelimits of point 0 and point 6.

FIG. 3 shows an embodiment of the principles of FIGS. 2 and 2a. Theletters A, B, C, D and E of FIG. 2 are repeated in FIG. 3, and areplaced near the various circuits performing the specific functions shownin block diagram in FIG. 2. An audio signal voltage at terminal 12causes a current to flow through resistor 3 to junction point 5, where avolt-age is produced that is fed to grid 74 of vacuum tube 16. Anamplified version of the same voltage appears at the plate 17 of tube16, and is attenuated an adjustable amount by gain control 19 beforebeing fed to the cathode follower tube 20. The output of tube 20 passesthrough an adjustable low-pass filter consisting of variable resistor 22and capacitor 23, and is then fed to cathode follower tube 21. Theoutput of tube 21 passes through an adjustable high-pass filterconsisting of capacitor 24 and variable resistor 25, and is then fed tothe grid 32 of the first phase shift tube 26a. A resistor 30 is theplate load resistor of tube 26a, and a resistor 29 which is nominallyequal to resistor 30 is used as the principal part of the cathode loadof tube 26a. Under these conditions, approximately equal voltages willappear at the plate 33 and at the cathode 34 of tube 26a; however, suchvoltages will differ by degrees in phase. The circuitry mentioned abovein connection with tube 26a is commonly known as a split load phaseinverter in that equal but opposite voltages are produced at the plateand cathode of the tube in question. From plate 33 of tube 26a aresistor 27 and capacitor 28 are con nected in series with the far endof the capacitor 28 connecting to cathode 34 of the same tube 26a.

It will now be apparent to those skilled in the art that the voltage (toground) at junction point 75 of resistor 27 and capacitor 28 will have amagnitude approximately equal to the magnitude of the voltages at plate33 and cathode 34 of tube 26a, and a phase angle which leads the voltageon grid 32 of tube 26a by an angle between zero and 180 degrees,depending on'the frequency. At very low frequencies the voltage at point75 will lead the voltage on grid 32 by almost 180 degrees, and at veryhigh frequencies the voltage at point 75 will approach the voltage atgrid 32 in phase angle. This voltage at point 75 is fed on to the secondphase shift tube 26b, which is exactly the same in operation as thefirst phase shift tube 26a, with the exception that the plate loadresistor 30 of first phase shift tube 26a is now replaced in the secondphase shift tube 26b with a resistor 76 and a variable resistor 31connected in series. The purpose of this arrangement is to provide ameans by which the output voltages of the second phase shift tube 26b,which appear at point 77, may be adjusted for approximate equality ofamplitude throughout the frequency range of interest. This adjustment isprovided by variable resistor 31, Which may be adjusted to compensatefor inequalities in the plate and cathode loading of first phase shifttube 26a and also provides the proper plate load adjustment for secondphase shift tube 26b. Resistor 76 is included in the circuit in order toallow variable resistor 31 to have a smaller total resistance and so bemore easily adjusted. If desired, resistor 76 could be omitted and itsresistance incorporated in variable resistor 31. In addition, ifsatisfactory equality of voltage throughout the frequency range ofinterest can be obtained at points 75, 77, etc., it is unnecessary tohave an adjustable plate load in every other phase shift tubes platecircuit, as shown by variable resistors 31, but such adjustment may beprovided only if deemed necessary after the signal has passed throughany convenient number of phase shift tube circuits.

These phase shift tubes constitute the all-pass phase shift network (s)A of FIG. 2. Two phase shift tubes with associated circuitry willproduce one peak in the response of the entire unit, since each stagewill shift the phase a maximum of 180 degrees, so that two stages arenecessary to cause the point 7 of FIG. 2a to make one completerevolution around the large circular path. For this reason the first twostages of phase shift tubes of FIG. 3 are referred to as the first pair;succeeding similar cascaded pairs of phase shift tube stages arereferred to as inten mediate pairs, and are indicated only by dottedline 42 of FIG. 3 in order to simplify the drawing. Such intermediatepairs may be visualized as a cascaded series of phase shift tube stageswith each stage identical to the phase shift tube stages of the firstpair, except that the adjustable plate load 31 may be used only asnecessary, and where it is not used a fixed resistor 30 is used toreplace resistor 76 and variable resistor 31; The last pair of phaseshift stages is shown in FIG. 3, and is the same as the first andintermediate pairs. After the signal has traversed all of the phaseshift stages, the output of the last pair of phase shift stages appearsat point 79, and is connected to contact 36 of switch 35.

When the arm of switch 35 is on contact 36, the output of these all-passphase shift networks is fed to grid 38 of isolation tube 14, and theoutput of tube 14- passes through resistor 15 to point 5. The originalinput signal applied to input terminal 12 produced a voltage at point 5,but the signal from tube 14 after passing through resistor 15 will nowact to modify the voltage appearing at point so as to produce a voltageat point 5 which is, in general, different from the voltage that wouldappear at point 5 due to the input signal alone. If we consider anygiven frequency in the input signal at terminal 12, it is apparentthatthe output of tube 14 will contain this frequency since such output isderived wholly from the input signal itself which contains thisfrequency. Hence, at this particular frequency, the voltage at point 5derived from tube 14 will add vectorially to the voltage at point 5derived from input 12. Therefore, the total voltage at point 5 at thisparticular frequency may be greater or smaller than the voltage thatwould appear at point 5 due to the input signal from terminal 12 alone.Likewise, the phase of the total voltage at point 5 may either lead orlag the phase of the voltage that would appear at point 5 due to theinput signal from terminal 12 alone. If the input signal contains .amultiplicity of frequencies, each component frequency voltage will beacted upon by the circuit of FIG. 3 in the manner just described for theinput voltage of a particular frequency. Hence, it is evident that thedevice of FIG. 3 will have a transfer characteristic from input 12 topoint 5 which is depicted by FIG. 1, because, with a constant inputvoltage at terminal 12, the vector sum of the two components of voltageat point 5 will vary with frequency because of the constantly changingphase shift with frequency in the all-pass phase shift networks A thatthe fedback signal from tube 14 has passed through before being combinedwith the input signal at point 5. Thus, the device of FIG. 3 producesthe desired effect.

Certain variations in the circuit of FIG. 3 are possible. The output atterminal 13 is shown as taken from plate 17 of tube 16 through blockingcapacitor 18 instead of being taken directly from point 5 via dottedconnection '76). This is done in order to take advantage of the gain oftube 16, since the voltage at its plate 17 is an amplified version of.the voltage at point 5 which is directly connected to grid 74 of thistube 16. Another variation would consist of additional contacts similarto contact 73 on switch 35. Contact 73 is connected by dotted conductor72 to circuit point 78, so that when the arm of switch 35 is on contact73, the signal will traverse one less phase shift stage than when thearm of switch 35 is on contact 36 before said signal is fed to the grid38 of tube 14. By providing a multiplicity of contact points on switch35 similar to contact 73, and with each such similar contact pointconnected to a different circuit point similar to circuit points 79, 78,77, 75, etc., it is possible by adjustment of switch 35' to cause thesignal to pass through any desired number of phase shift stages beforebeing fed on to tube 14. This will then provide an adjustment of thenumber of transmission peaks introduced by the device, as well as anadjustment of the total phase change through out the audio frequencyspectrum produced by the circuit between input terminal 12 and outputterminal 13. It should be noted that the gain control 19 will alsoprovide an adjustment of the total phase change produced by the device,but will not affect the number of transmission peaks, although it willprovide an adjustment of the height of these peaks. In general, it wouldseem desirable to have as many pairs of phase shift stages included inthe intermediate pairs at 42 as possible or economically practical.Experiments with the electronic filter of FIG. 3 have indicated that itis generally necessary to have more than eight peaks in the response vs.frequency curve if the beneficial effects of this basic method of tonequality improvement are to be achieved to any practical degree.

'The foregoing discussion should now make apparent the fact that thedevice disclosed in FIGS. 2 and 3 will simulate substantially the actionof the rigid parts of the violin, as described, or the air column in awind instrument, etc. No attempt is made herein to simulate exactly thetonal quality of any particular orchestral instrurnent by the use of thesubject device. That would require a selection of the proper number ofintermediate pairs of phase shift circuits, and a selection of theproper values of the capacitors 28 and resistors 27 of FIG. 3, togetherwith proper adjustment of the gain control and equalizer to approximatethe variation of transmission with frequency which is characteristic ofany specific orchestral instrument.

When the input signal to terminal 12 is derived from a tone generatorhaving high frequency stability, as found in most electrical andelectronic organs, it should be noted that it is necessary to use'thevibrato built into the electronic organ in order to provide thenecessary frequency instability to allow the production of the beatsreferred to, since the tone generators used in electronic organs aremuch too stable in frequency to allow the vary phase shift vs. frequencycharacteristic of any device following the basic method of FIG. 1 toproduce its beneficial effect. It should also be noted that theintroduction of the large number of peaks in the response vs. frequencycurve causes musical tones separated by as little as a semitone to havesignificantly different harmonic structures, and there will beconsiderable variety in the tonal quality of the different notes of thescale. This effect, when not over done, is usually consideredbeneficial; in fact, many of the instruments of the orchestra exhibitthe same changing tonal quality as they play the musical scale. Sucheffect is to be expected because of the action of the conventionalmusical instruments in altering the harmonic structure of the waveproduced at the point of generation, such as the string of a violin vorthe mouthpiece of a trumpet, etc. This effect is one more indication ofthe fact that the overall action of any device following the basicmethod of FIG. 1 is the counterpart of the action of the body of thevarious orchestral instruments or of the air column of the windinstruments as applied to electric and electronic musical instruments.It is, however, possible to limit the differences in the harmonicstructure of the different notes of the scale by limiting the ratio ofeach maximum transmission to the next adjacent minimum transmissionwhile retaining the desired total phase change throughout the audiorange by having a sufficient number of transmission peaks in this audiorange. it should be emphasized that, to produce the maximum beatingeffect, as described, it is advantageous to use as many transmissionpeaks as possible, since the total phase change will thereby increaseproportionally so long as the peak to valley ratio is kept constant.

It would not be possible to use only the all-pass phase shift network(s)A of FIG. 2 as a device to produce the desirable beating effect that hasbeen described, since such an attempt would result in the phase shiftsof the various harmonics of a signal being all in the same direction, sothat very little beating would result. However, the allpass phase shiftnetwork(s) A could be used alone to produce considerable beating effectif the output were taken from a multiplicity of points instead of from asingle point of the circuit. For example, the phase shift stages aloneof FIG. 3 could be used in this manner if, with circuit parts B, C, Dand E deleted, the input terminal 12 were to connect to grid 32 of firstphase shift tube 26a, and the output terminal 13 were to be connectedthrough suitable isolation resistors to two or more of, for example, thecathodes 34 of the phase shift tubes. Such an arrangement, however,would be difficult to adjust and control, and would, in general, producean amplitude vs. frequency characteristic having apparently erraticmaxima and minima, and a phase shift vs. frequency characteristicsomewhat less desirable than that produced by the unmodified circuit ofFIG. 3.

If the signal input to any device following the basic method of FIG. 1has little or no instability in its frequency, the subject devices willproduce little or no ulti mate beating effect, as described. To overcomethis difficulty in another manner, a vibrato may be built into thesubject tone color devices. An embodiment of such a vibrato addition,suitable for use with the circuit of FIG. 3, is shown in FIG. 3a.

In FIG. 2 is also shown a symbolic representation of the effect of thecircuit of FIG. 3a. The vibrato addition is added to the circuit of FIG.3 when the moving arm of switch 35 shown in FIGS. 2 and 3 is thrown toposition 37. In FIG. 2, this causes the input to B to come from themoving arm 43 of the symbolic potentiometer 44, the ends of which areconnected to the input and output of allpass phase shift network(s) A.As the vibrato addition causes the moving arm 43 of this symbolicpotentiometer 44 to move from the one end to the other end, the input toB will alternately be derived principally from the input to A and theoutput of A. This action alternately removes and then reinserts thephase shift network(s) A in the system shown.

In FIG. 3a is shown a circuit including one tube 45 as vibrato frequencyoscillator (frequency range approximately 4 to cycles per second); asecond tube 46 is used as a phase splitter to produce two voltages ofequal amplitude but 180 degrees apart in phase; this phase splitter 46is fed from the oscillator tube 45 and so produces an output of vibratofrequency. These two vibrato frequency signals 180 degrees apart inphase are then fed to the third tube 47 which is a double triode havinga common plate load resistor 48 for both triodes. The two vibratofrequency signals are fed to the two grids 51 and 52 of this tube 47,one signal to each grid, via capacitors 49 and resistors 50. To thesesame two grids 51 and 52 are fed signals from cathodes 34 (of firstphase shift tube 26a) and 34a (of last phase shift tube 26b) of FIG. 3.The signal to grid 51 comes from cathode 34 via conductor 39, and thesignal to grid 52 comes from cathode 34a via conductor 40. Suitableresistive voltage dividers comprising resistors 50, 53, 54, 55 and 56are used to equalize these two signal voltages on the llLWO grids 51 and52. This third tube 47 will now act .as a mixer tube under the controlof the vibrato frequency signals to produce in its plate circuit 57 asignal having a periodic variation in the ratio of the signal derivedfrom cathode 34 to the signal derived from cathode 34a. This effect isproduced by the opposite changes of transconductance of the two halvesof this mixer tube 47 produced by the variations in the grid voltagescaused by the vibrato frequency signals. The mixed signal now appearingacross the common plate load resistor 48 of the two triodes of thismixer tube 47 is now passed through a high-pass filter 58 to remove mostof the vibrato frequency signal and its lower harmonics, and is then connected to contact 37 of the switch 35 of FIG. 3 via conductor 41, and(when the arm of the switch 35 is moved to contact point 37) is fed backto grid 38 of tube 14 in the circuit shown in FIG. 3.

The action of the three additional tubes 45, 46 and 47 may now besummarized as follows: The vibrato frequency oscillator 45, the phasesplitter 46, and the mixer tube 47 result in a periodic variation in theeffect of the phase shift circuits of FIG. 3 (A of FIG. 2) at a vibratofrequency rate; that is, the circuit of FIG. 3 is caused to act so thatthe signal appearing at the arm of switch 35 is derived mostly from thefirst phase shift tube cathode 34 when the vibrato oscillator 45 isproducing an output of a certain polarity, and, when the vibratooscillator 45 is producing an output of the opposite polarity, thesignal appearing at the arm of switch 35 is derived mostly from cathode34a. Since the phase shift of the entire instrument being described,from input to output (at any particular frequency), is a function of thenumber of phase shift stages (the pairs of FIG. 3) that the signalpasses through before being fed back to the circuits connected to thearm of switch 35, it may be seen that the phase shift from input tooutput of the entire instrument will vary at the vibrato frequency rate,and the output of the instrument will then be frequency modulated, or,in common musical terminology, a vibrato has been introduced into thesignal.

The use of a circuit such as shown in FIG. 3a in conjunction with thecircuit of FIG. 3 is to enable the instrument to produce an outputsignal capable of exciting an auditory sensation of beating, aspreviously referred to, even when the input signal has no frequencyvariation whatever. Any given harmonic component of the input signalwill then emerge from the output of the instrument with its frequencyvarying back and forth at the vibrato rate, as just described. However,since each harmonic is a different frequency, each will be affected in adifferent manner by the action just described. That is, the frequencyvariation of the different harmonics will be of varying amount, and, ingeneral, of varying sign; some harmonics will be increasing in frequencyat various rates at the same instant that other harmonics are decreasingin frequency at various rates. These harmonics will ultimately produce apleasing auditory effect similar to that described. To produce a stillgreater effect of this type, it is possible to use this vibrato additionwith the input signal already frequency modulated by a vibrato signalfrom some other source.

FIG. 4 is a block diagram of another embodiment of the basic method ofFIG. 1. To obtain the Desired Output Voltage from terminals 59, anelectroacoustic or electromechanical transducer 60 is coupled to anacoustic or mechanical device 61. The transducer 6t may be any of thecommon types such as the electrodynamic (moving conductor),electrostatic, magnetic, magnetostriction, or piezoelectric types. Theacoustic or mechanical device 61 may have an almost infinite variety offorms; typical examples of acoustic devices would be the air or othergas or liquid column or cavity inside a tube or horn or other cavity ofalmost any conceivable shape, with or without openings to the outsideair. Similar examples of mechanical devices would be rods, tubes, bells,boxes, solid resonators with or without internal cavities which may ormay not be accessible from the outside, such resonators having almostany conceivable shape. These acoustic or mechanical devices 61 aredesigned to have a multiplicity of resonances within the audio frequencyrange,

impedance-s, and impedance 86 passive.

such resonances being preferably not harmonically related. The effect ofthese resonances is reflected back into the electrical circuit toproduce a variation in the electrical impedance of the transducer 60. Ifa Controlled Current as represented at 62 is passed through the circuitcontaining the transducer 60, the voltage appearing across the circuitat output terminals 59 may be made to exhibit as many or almost as manypeaks in its responsive vs. frequency curve as there are minima ofacoustic or mechanical impedance in the device 61 coupled to thetransducer 60. This effect of transferring acoustic or mechanicalimpedances and variations of the same back into an electrical circuit isset forth very clearly in the book by F. V. Hunt,Electroacoustics"-Cl1apter 2 in particular. The Controlled Current 62(the input signal) may be controlled by controlling the source impedancefrom which it is derived, and/ or by controlling its magnitude withrespect to frequency. Network 63 is used to control and modify theoverall effect of the entire device in producing response peaks andphase shifits; it could be omitted if the effect of the device weresatisfactory without it. If network 63 is composed only of passivecomponents, the power source 64 is not needed; however, if network 63contains active components (vacuum tubes, transistors or otheramplifying devices), then power source 64- will be needed to supply therequired operating potentials of these active components. Although thedescriptions of the diagrams showing some of the preferred embodimentsof the present invention may mention the production of peaks in theresponse vs. frequency curve more frequently than the production ofphase shift changes vs. frequency, it should be remembered that this isdone only because of the convenience of visualizing the action of suchdevices from this standpoint; the phase shift changes which necessarilyaccompany these peaks in the response constitute the basic and primaryreason for the use of these devices in connection with electronic orelectric musical instruments.

FIG. 4a shows a possible embodiment of the block diagram of FIG. 4. Themulti-resonant device is the air column 80' contained in tube 82 andcavity 81. The electroacoustic transducer is the electrodynamic earphone83, which is tightly coupled to the air column 88 through the onlyopening in the tube. and cavity assembly 82 and 81. The transducer 83 isconnected to the active network composed of impedances 84, 85 and 86arranged in a T configuration. This T configuration is particularlyuseful with transducers functioning by the action of magnetic fields,such as the magnetic, electrodynamic (moving conductor) andmagnetostriction types. In the embodiment of FIG. 4a, the impedances 84and 85 are active The active impedances are used in order to neutralizeor partially neutralize the effect of the electrical and mechanicalparameters of the transdcer 83 which would otherwise tend to obscure theimpedance changes of transducer 83. In particular, impedance 84 is thenegative of the blocked impedance of the transducer 83, and impedance 85is the negative of the impedance of the shunt portion of the equivalentelectrical circuit of transducer 83, which shunt portion isrepresentative of the reflected electrical effect of the entire movingcoil assembly of transducer 83. Impedance 85 is a resistor which acts tolimit the minimum values of impedance offered to the controlled current87. If it were desirable to limit these minimum impedances more at thelower audio frequencies than at the higher frequencies, then. impedance86 could be in the form of a resistor shunted by a capacitor; if it weredesirable to limit the minima of impedance more at the higher than atthe lower audio frequencies, then impedance 86 could be in the form of aresistor in series with an inductance. Many variations in the impedances84, 85 and 86 are possible, so that it is possible to satisfactorilycontrol the impedance presented to the controlled current 87 at audiofrequencies.

The general action of the device of FIG. 4a is a follows: The air column88 will exhibit acoustic impedance variations at the opening to whichtransducer 83 is fastened. A minima of acoustic impedance will bepresented to the transducer 83 at the lowest resonant frequency of theair column 88; as the frequency is increased, the acoustic impedancewill increase until a maximum is reached, and, as the frequency isincreased still more the acoustic impedance will fall again until asecond minimum is reached. This process will repeat itself as thefrequency is increased still further, so that successive maxima andminima of acoustic impedance will be presented to the transducer 83.Each minima of acoustic impedance will cause a maxima of electricimpedance to appear at the terminals of transducer 83, and each maximaof acoustic impedance will cause a minima of electric impedance. Theseelectrical impedance variations may, however, be rather small because ofthe obscuring effect of the electrical and mechanical parameters oftransducer 83. and 85 will act, as described, to increase theseelectrical impedance variations, and impedance 86 will modify them asdesired. Since the impedance which is presented to the controlledcurrent 87 has now been made to vary in the desired manner, it isobvious from the alternating current form of Ohms Law that the voltagedeveloped across this impedance and fed to output terminals 88 will beproportional to this impedance if the controlled current is derived froma very high impedance source. Hence as the impedance of the device goesthrough successive maxima and minima, the desired output voltage atterminals 88 will likewise go through successive maxima and minima, anda transfer function such as is shown in FIG. 1 is obtained, as desired.It is, of course, obvious that either or both of the active impedanceelements 84 and 85 may be omitted or replaced with passive impedances ifsatisfactory results are obtained in this manner. The large variationsthat may be found in the types of air column (length, diameter, etc.)and in the transducer 83 make it difficult .to generalize concerning theexact characteristics of each of the impedances 84, and 86; the mostappropriate values must be determined in each individual case. Althoughtransducer 83 was specified as being an electrodynamic earphone, it isobvious that any type of electro-acoustic transducer could also be usedprovided the other components associated with it were suitably chosen tooperate in a satisfactory manner in conjunction with the variouscharacteristics of the particular transducer used. The diameter of theair column 80 must be selected to provide a suitable load for the movingelement of transducer 83; similarly, the length of air column 80 and thesize of cavity 81 must be selected to provide the most suitabledistribution of resonances throughout the audio frequency range. Inorder to conserve space, tubing 82 may be coiled as desired.

The use of cavity 81 may be explained as follows: If the cavity 81 wereomitted and the right hand end of tube 82 closed, the frequencies ofminimum acoustic impedance presented to the transducer 83 would beproportional to the following integers: 1, 3, 5, 7, 9, 11, 13, etc. Itmay be seen that the number of these resonances in any given octavecontaining such resonances, will, in general, be more than the number ofsuch resonances in the next lower octave. Since a signal containing allharmonics of the fundamental frequency also has more harmonics in anygiven octave than in the next lower octave, we see that the action ofthe air column tends to fit the signal, in that more phase undulationsper octave are available in the upper octaves where the signalsharmonics are most numerous. However, it is not desirable that theaction of the air column fit the signal exactly, even if it be at onlyone signal frequency. If this were to occur, then the fundamentalfrequency of the signal would occur at the lowest frequency of minimumacoustic impedance, the third harmonic of the signal would be at thenext higher frequency of minimum Impedances 84 13 acoustic impedance,the fifth harmonic of the signal would be at the third minimum ofacoustic impedance, and so on. Unless the amplitude peaks in theelectrical response of the device were small, the occurrence of such acondition would produce a disproportionately large increase in theamplitude of the signal at this particular frequency. To prevent such anoccurrence the cavity 81 has been used. The use of an acousticalcapacity such as cavity 81 will prevent the minima of acoustic impedancefrom following the 1, 3, 5, etc. series given above, so that thecondition would not occur where the fundamental frequency of the signalwere the same as the lowest frequency of the series.

Devices of electroacoustic or electromechanical nature would possessboth cost and weight advantages over the electronic filter of FIG. 3,and at the same time could be made to produce better results in thattheir resonances tend to fit the signal better than the resonances ofthe electronic filter of FIG. 3 could economically be made to do.

FIG. is an embodiment of the present invention which is very similar toFIG. 4. Multi-resonant device 89 may be identical to 61; transducer 90may be the same as 60; and network 91 performs the same function as 63.FIG. 5, however, instead of being fed from a high impedance currentsource is fed from a low impedance voltage source. The impedanceundulations at the input to network 91 will cause corresponding changesin the current taken from the voltage source connected to inputterminals 93. This changing current flows through resistor 92, which hasa resistance value sufliciently low to not interfere greatly with thecurrent changes produced by components 89, 90 and 91. An output voltagewill therefore be produced across resistor 92 and appear at outputterminals 94. This voltage will have the same undulating characteristicwith respect to frequency as the output voltage of the system of FIG. 4.The system of FIG. 5 differs from that of FIG. 4, however, in that it isnow possible to adjust the source impedance of the controlled inputvoltage to terminals 93 and the value of resistor 92, together with thecomponents in network 91 so that all the electrical parameters togetherreflect an impedance into the mechanoacoustic system composed of themoving elements of transducer 90 and multi-resonant device 89 of such amagnitude that the multi'resonant device 89 is connected to itscharacteristic acoustic or mechanical impedance at the point where thetransducer 90 is coupled to it. This arrangement has the advantage thata minimum of what is variously called reverberation, hangover, orringing is transmitted to the output terminals 94 after the removal of avoltage applied to input terminals 93. The desirability of a devicefollowing FIG. 5 compared to that following FIG. 4 must be decided bythe preference of the listener.

A possible embodiment of FIG. 5 is shown in FIG. 5a. An air column 95 isformed by the air in tubes 97 and 96, which two tubes have differentinternal diameters, and are securely joined together, with theright-hand end of tube '96 closed, and with the left-hand end of tube 97tightly coupled to electrostatic transducer 98 as shown. The electricalconnections to transducer 93 are connected to a pi network composed toimpedances 99, 100 and 101. Such a pi network is useful with transducersdepending on the electric field for their operation, such as thepiezoelectric and electrostatic types. In the embodiment of FIG. 5a, theimpedances 99 and 100 are active impedances, and impedance 101 passive.The active impedances are used in order to neutralize or partiallyneutralize the effects of the electrical and mechanical parameters ofthe transducer 98 which would otherwise tend to obscure the impedancechanges of transducer 98. In particular, impedance 99 is the negative ofthe static electrical capacitance of electrostatic transducer 98, andimpedance 100 is the negative of the impedance of the series portion ofthe equivalent electrical circuit of transducer 98, which series portionis principally representative of the reflected electrical eflfect of themechanical constants of the diaphragm of transducer 98. Impedance 101 isa resistor which acts to limit the maximum values of impedance insertedin the circuit by transducer 98 acting with active impedances 99 and100. Many variations in the impedances 99, 100 and 101 are possible, sothat it is possible to satisfactorily control the impedance inserted inthe circuit throughout the audio frequency range. In general, theimpedances 99, 100 and 101 have a function analogous to that ofimpedances 84-, 85 and 86 of FIG. 4a, so that the same or equivalentcomments apply to both sets of impedances.

The general action of the embodiment of FIG. 5a is analogous to that ofFIG. 4a; however, in FIG. 5a the maxima of acoustic impedance will bereflected into the electrical circuit as maxima of electrical impedance,and vice versa. Since the impedance which has been introduced into thecircuit by air column 95, electrostatic transducer 98, and impedances99, and 101 has been made, by suitable adjustment of these parameters,to exhibit a multiplicity of successive maxima and minima throughout theaudio range, it is obvious that the current passing through resistor103, and hence the output voltage appearing at terminals 104 willlikewise exhibit a similar multiplicity of maxima and minima. Withrespect to the selection of air column 95, tubing 97 and transducer 98,the previous comments concerning the corresponding components of FIG. 4aare generally applicable. Tubing 96, having a different internaldiameter than tubing 97, is used to accomplish the same purpose as iscavity 81 of FIG. 4a. For greatest efliciency, the input voltage sourceconnected to terminals 102 should have as low an internal impedance aspossible, so that the resistor 103 may be as large as possible in orderto produce the maximum output signal at terminals 104. There will,however, in any given setup embodying the various components shown inFIG. 5a, be some one particular value of resistor 103 which will cause aminimum of reverberation as discussed in connection with FIG. 5.

FIG. 6 shows another block diagram embodiment of the basic method ofFIG. 1. The signal is applied to input terminals 105 which are directlyconnected to electroacoustic or electromechanical transducer 106, whichis coupled to multi-resonant acoustic or mechanical device 107. Alsocoupled to resonant device 107 is another electroacoustic orelectromechanical transducer 108, the electrical output of which isconnected to output terminals 109. Transducers 106 and 103 and resonantdevice 107 may be the same as any of the corresponding devices mentionedin connection with FIG. 4. The mode of operation embodied in the blockdiagram of FIG. 6 does not depend upon the principle of reflectedimpedances, but is a type of operation similar to the action of therigid parts of a violin (for example) or the open air column of a windinstrument in that the resonances are introduced by a transmission ofthe signal through the multi-resonant device 107 from input transducer106 to output transducer 108.

A specific embodiment of the principle of FIG. 6 is shown in FIG. 6a,which illustrates one possible form such a device might take in which itis desired to imitate the resonance pattern of a hollow-bodied stringinstrument such as a violin. Signal input to terminals 110 is applied tocoil 111 which is wound on nickel rod 112. Coil 111 and rod 112 comprisea magnetostrictive transducer which is tightly coupled to wooden shell113. Microphone 114 picks up the vibrations in shell 113 and produces anoutput voltage which appears at terminals 116. As the signal inputfrequency is varied, the output from terminals 116 will exhibit variouspeaks in its response curve, depending on the various resonances ofshell 113. Variations in the placement of nickel rod 112 and microphone114 will cause considerable variety in the response pattern that isproduced. Sound proof box 115 is used to prevent basic method of FIG. 1.

components may be of the same type as those described in connection withFIG. 6. In FIG. 7, however, resonant device 119 is allowed to radiateacoustic energy 1% directly to the surrounding air. The signal appliedto input terminals 117 must have sufficient power capability to producea useful acoustic signal. Since multi-resonent device 119 will radiatemost efficiently at its resonant frequencies, an undulating response vs.frequency will be obtained as desired. FIG. 6a will serve to illustratea specific embodiment of FIG. 7 if sound-proof box 115, microphone 114,and output terminals 116 are all omitted, so that resonant shell 113 maybe allowed to radiate sound directly to the atmosphere. It is obviousthat the method of FIG. 7 most closely simulates the effect of an actualinstrument of the orchestra, the only diiference being that thevibrations are transmitted to the multiresonant device by the use of atransducer, whereas with an orchestral instrument the vibrations are adirect result of the physical effort of the musician.

FIG. 8 illustrates the use of lumped electrical constants to satisfy thebasic method of FIG. 1. In this embodiment, a controlled current 121 isapplied to terminal-s 122, which terminals serve both as current inputterminals and voltage output terminals. Current 121 may flow through amultiplicity of series resonant circuits all connected in parallel. Thefirst, which is composed of inductance 123, resistor 124 and capacitor125, is tuned to some particular frequency; the second, which iscomposed of inductance 126, resistor 127 and capacitor 128,

will be tuned to a different frequency; other similar circuitssymbolized by dashed lines 129 will each be tuned to its own particularresonant frequency; and the final series circuit, which is composed ofinductance 130, resistor 131 and capacitor 132, will be tuned to thelast resonant frequency that we desire to include in this embodiment.Although resistors 124, 127 and 131 may be actual physical devices, itwill more frequently be found that it is desirable to reduce theresistance of each series circuit to a minimum, so that these resistorswill then be symbolic of the total effective residual resistance of eachcircuit. As is well known to those skilled in the art, in accordancewith Fosters Reactance Theorem (Bell Sys. Tech. Journ., April 1924), thevoltage appearing across the terminals 122 of FIG. 8, When fed from asuitable current source such as a constant current source, will exhibitcharacteristics such as illustrated in FIG. 1. It is possible to arrangethe components of FIG. 8 in various ways to produce the same orequivalent results; several such rearrangements are shown in Chap. V ofErnest A. Guillemins book Communication Networks, vol. 11. Still othermodifications are possible, such as the use of one or more resonantcircuits in the circuitry of each stage of successive stages of vacuumtube amplification. Each of such cascaded stages need not necessarilycontain inductive elements, but may be of the resistancecapacitancefeedback type, such as the parallel-T or bridged-T frequency-selectivefeedback amplifier or some -modification or equivalent thereof.

All of the various embodiments of the present invention which have nointernal vibrato (such as the vibrato addition of FIG. 3a supplies tothe embodiment of FIG. 3) need some type of frequency instabilitypresent on the input signal if the effectiveness of the device is to berealized. However, if no such frequency instability is present or if itis desired to incorporate an internal vibrato in any of the embodimentsof the present invention, a suitable vibrato addition can generally beadded to the various embodiments. Such a vibrato addition would consistof a device to periodically vary the resonant frequencies of theacoustic or mechanical device coupled to the transducer(s) at a vibratofrequency rate. For example, a vibrato addition could be added to theembodiment of FIG. 4a by the use of additional transducers, as shown inFIG. 4b, designed to operate in the vibrato frequency range (about 4 to10 cycles per second). Trans ducer 133 is connected by wires 135 to asuitable source of vibrato-frequency voltage, and transducer 134 isconnected by wires 136 to the same source of voltage, but in such mannerthat as the diaphragm of transducer 133 is driven to the left (in FIG.4b), the diaphragm of transducer 134 is driven upward, and vice versa.Hence, if the transducers 133 and 134 have similar characteristics, thepressure in air column will not be appreciably altered by the combinedmotions of their diaphagms. However, many of the resonant frequencies ofair column 30 will be altered by the action of transducers 133 and 134at the vibrato frequency rate, thus producing a vibrato-frequencyvariation of the impedance reflected into the electrical circuits oftransducer 83, as desired. These additional transducers and associatedcircuitry would be designed to present a high acoustic impedance tonormal audio frequencies, and would therefore not greatly interfere withthe resonance pattern at any particular instant, even though they wouldbe varying this pattern from instant to instant. Other devices toproduce the same or equivalent results could readily be devised by thoseskilled in the art for all embodiments of the present invention.

It is, of course, to be understood that wherever thermionic vacuum tubesare mentioned in this disclosure, the same or equivalent results couldbe achieved by the use of transistors or other solid-state amplifyingdevices. I consider the alternate use of such devices to be entirelywithin the scope of the present invention.

The term feedback or coupling, as used in the claims, will be understoodto include not onlyelectronic feedback but also feedback or couplingthrough mechanical means as well, in accordance with this disclosure.

Since the present invention will probably find its greatest applicationto electric or electronic organs, it might be,

-to such organs. The signal which normally went to the input of thepower amplifier of the organ could be fedinstead to the input of one ofthe embodiments of the present invention. The output of this embodimentcould then be fed to the input of the power amplifier of the organ.However, it might be desirable to apply the present invention to onlyone manual of the organ at a time; in this case the output signal of thedesired manual, after passing through all of the tone filters andcouplers connected with that manual, would be fed to the input of anembodiment of this invention, the output of which would be fed to themixer or to the separate power am-- plifier which might be associatedwith that particular manual. Again, it might be desirable to have oneembodiment of this invention for each manual of the organ, with each ofthe various embodiments having different resonance patterns. Othermethods of applying this invention to electrical organs or otherelectronic musical instruments will be readily apparent to those skilledin the art.

Although the present invention has been disclosed in connection withcertain preferred embodiments thereof, it will be apparent to thoseskilled in the art that many modifications and variations thereof may bemade without departing from the fundamental principles of the invention.I therefore desire by the following claims to include within the scopeof my invention all such variations and modifications by whichsubstantially the results of the invention may be obtained by the use ofsubstantially the same or equivalent means.

This is a divisional application of United States application Serial No.537,134, filed September 28, 1955, for Device for Altering Tone Qualityin Electronic Musical Instrument or the Like.

I claim:

1. A device for producing a modified vibrato in an electronic musicalinstrument or the like comprising an input circuit, an output circuit, amultiresonant acoustic device, and transducer means including at leastone transducer coupling said multiresonant device to said input andoutput circuits to produce more than eight peaks in the transferresponse versus frequency of the signals in said output circuit withrespect to the signals in said input circuit.

2. A device according to claim 1, wherein the multiresonant acousticmeans propagates sound Waves.

3. The device according to claim 1 in which the said input circuit hasat least one input terminal to which signals having a vibrato frequencycharacteristic are applied, and in which the said output circuit has atleast one output terminal at which a modified phase shifted signal isproduced, and means coupled to said at least one output terminal toprovide an audible output having the desired phase shift changes.

4. The device according to claim 1 in which said transducer means incombination with said multiresonant device produces an effectiveelectrical impedance and phase angle which varies with frequency.

5. The device according to claim 4 in which a voltage is produced acrossthe effective impedance of said transducer means and multiresonantdevice in combination in response to a controlled signal current in saidinput circuit, said voltage being applied in said output circuit toproduce an output signal having the desired multiplicity of peaks in itsresponse versus frequency.

6. The device according to claim 5 in which said multiresonant device isan acoustic pipe-like structure.

7. The device according to claim 4 in which the effective impedance ofsaid transducer means and multiresonant device in combination modifiesthe current flow produced by a controlled signal voltage in said inputcircuit, said current acting in said output circuit to produce an outputsignal having the desired multiplicity of peaks in its response versusfrequency.

8. The device according to claim 7 in which said multiresonant device isan acoustic pipe-like structure.

9. The device according to claim 1 in which the transducer meanscoupling said multiresonant device includes a first transducercoupled'to said input circuit and a second transducer coupled to saidoutput circuit.

10. The device according to claim 9 in which said multiresonant deviceis a hollow vibratory body.

11. The device according to claim 9 in which said multiresonant deviceis an acoustic structure.

12. Apparatus for producing a modified vibrato in an electronic musicalinstrument or the like comprising input terminals, output terminals, amultiresonant acoustic device, and transducer means including at leastone trans ducer coupling said multiresonant device to said input andoutput terminals to produce more than eight peaks in the response versusfrequency characteristics of the apparatus.

13. The device according to claim 12 in which signals having a vibratofrequency characteristic are applied to the said input terminals, and inwhich a modified phase shifted signal is produced at the said outputterminals, and means coupled to said output terminals to provide anaudible output having the desired phase shift changes.

14. The device according to claim 12 in which said transducer means incombination with said multiresonant device producesan effectiveelectrical impedance and phase angle which varies with frequency.

15. The device according to claim 14 in which a voltage is producedacross the effective impedance of said transducer means andmultiresonant device in response to a controlled signal current appliedto said input terminals, said voltage being coupled to said outputterminals to produce an output signal having the desired multiplicity ofpeaks in its response versus frequency.

16. The device according to claim 15 in which said multiresonant deviceis an acoustic pipe-like structure.

17. The device according to claim 14 in which said transducer means andmultiresonant device modifies the current flow produced by a controlledsignal voltage applied to said input terminals, said current flowproducing the desired modified version of the input signal voltageacross said output terminals.

18. The device according to claim 17 in which said multiresonant deviceis an acoustic pipe-like structure.

19. The device according to claim 12 in which the transducer meanscoupling said multiresonant device includes a first transducer coupledto said input terminals, and a second transducer coupled to said outputterminals.

20. The device according to claim 19 in which said multiresonant deviceis a hollow vibratory body.

21. The device according to claim 19 in which said multiresonant deviceis an acoustic structure.

References Cited in the file of this patent UNITED STATES PATENTS1,921,501 Bower Aug. 8, 1933 2,230,836 Hammond Feb. 14, 1941 3,031,909White May 1, 1962

1. A DEVICE FOR PRODUCING A MODIFIED VIBRATO IN AN ELECTRONIC MUSICALINSTRUMENT OR THE LIKE COMPRISING AN INPUT CIRCUIT, AN OUTPUT CIRCUIT, AMULTIRESONANT ACOUSTIC DEVICE, AND TRANSDUCER MEANS INCLUDING AT LEASTONE TRANSDUCER COUPLING SAID MULTIRESONANT DEVICE TO SAID INPUT ANDOUTPUT CIRCUITS TO PRODUCE MORE THAN EIGHT PEAKS IN THE TRANSFERRESPONSE VERSUS FREQUENCY OF THE SIGNALS IN SAID OUTPUT CIRCUIT WITHRESPECT TO THE SIGNALS IN SAID INPUT CIRCUIT.